1. Field of the Invention
The present invention relates to a control circuit and, more particularly, to a motor control circuit for controlling an H-bridge drive circuit to operate a bi-directional motor among a forward rotation mode, a brake mode, and a, reverse rotation mode.
2. Description of the Related Art
FIG. 1(a) is a detailed circuit diagram showing a conventional motor control circuit 10. FIG. 1(b) is a timing chart showing an operation of the conventional motor control circuit 10. Referring to FIG. 1(a), a bi-directional motor M is connected with an H-bridge drive circuit 1, and the motor control circuit 10 operates the bi-directional motor M among a standby mode, a forward rotation mode, a brake mode, and a reverse rotation mode through controlling the H-bridge drive circuit 1. More specifically, the H-bridge drive circuit 1 is constructed by six drive transistors Q1 to Q6. A terminal Na of the bi-directional motor M is coupled to a collector electrode of the pnp transistor Q1 and a collector electrode of the npn transistor Q3. A terminal Nb of the bi-directional motor M is coupled to a collector electrode of the pnp transistor Q2 and a collector electrode of the npn transistor Q4. Emitter electrodes of the transistors Q1 and Q2 are connected together to a drive voltage source Vd while emitter electrodes of the transistors Q3 and Q4 are connected together to a ground potential. The transistors Q1 to Q4 are provided with flywheel diodes D1 to D4, respectively, for allowing the electromotive force-induced current to flow through when the transistors Q1 to Q4 are turned off. The npn transistor Q5 has a collector electrode coupled to a base electrode of the transistor Q1, and an emitter electrode coupled to a base electrode of the transistor Q4. The npn transistor Q6 has a collector electrode coupled to a base electrode of the transistor Q2 and an emitter electrode coupled to a base electrode of the transistor Q3.
The motor control circuit 10 has a forward rotation current source constructed by a pnp transistor Q7 and a reverse rotation current source constructed by a pnp transistor Q8. The transistor Q7 is biased by a control voltage source Vc1 for providing a control current Ic1. The transistor Q8 is biased by a control voltage source Vc2 for providing a control current Ic2. The control current Ic1 is split at a diverging node N1 into a first partial current If1 and a second partial current If2. The first partial current If1 is supplied to the base electrode of the transistor Q5 while the second partial current If2 is supplied to the base electrode of the transistor Q4. On the other hand, the control current Ic2 is split at a diverging node N2 into a first partial current Ir1 and a second partial current Ir2. The first partial current Ir1 is supplied to the base electrode of the transistor Q6 while the second partial current Ir2 is supplied to the base electrode of the transistor Q3.
In response to a forward rotation command signal FWD and a reverse rotation command signal REV, the motor control circuit 10 controls the motor drive circuit 1 to operate the motor M among a standby mode, a forward rotation mode, a brake mode, and a reverse rotation mode. Each of the forward and reverse rotation command signals FWD and REV is a logic signal having a HIGH state and a LOW state. When the forward rotation command signal FWD and the reverse rotation command signal REV are both at the LOW state, npn transistors Q11 and Q12 are turned off. As a result, the control currents Ic1 and Ic2 supplied from the control voltage sources Vc1 and Vc2 are zero, which may be exemplified by periods T1, T4, and T7 shown in FIG. 1(b). In this case, the motor control circuit 10 is prohibited from supplying the four partial currents If1, If2, Ir1, and Ir2 and then makes all of the drive transistors Q1 to Q6 nonconductive, thereby operating the motor M in the standby mode.
When the forward rotation command signal FWD is at the HIGH state and the reverse rotation command signal REV is at the LOW state, the transistor Q11 is turned on and the transistor Q12 is turned off. The control current Ic1 supplied from the control voltage source Vc1 has a maximum Ic1m and the control current Ic2 supplied from the control voltage source Vc2 is zero, which may be exemplified by period T2 shown in FIG. 1(b). At this time, the first partial current If1 of the control current Ic1 is allowed to be supplied to the base electrode of the drive transistor Q5 because npn transistor Q9 which is controlled by the reverse rotation command signal REV is turned off. As a result, the transistors Q1 and Q4 are turned on for allowing a forward current to flow from the terminal Na to the terminal Nb, thereby operating the motor M in the forward rotation mode.
When the forward rotation command signal FWD and the reverse rotation command signal REV are both at the HIGH state, the transistors Q11 and Q12 are turned on. The control current Ic1 supplied from the control voltage source Vc1 has a maximum Ic1m and the control current Ic2 supplied from the control voltage source Vc2 has a maximum Ic2m, which may be exemplified by periods T3 and T6 shown in FIG. 1(b). At this time, the first partial current If1 of the control current Ic1 is bypassed down to the ground potential though the transistor Q9 which is also turned on by the reverse rotation command signal REV, and therefore prevented from being supplied to the base electrode of the drive transistor Q5. Likely, the first partial current Ir1 of the control current Ic2 is bypassed down to the ground potential though the transistor Q10 which is also turned on by the forward rotation command signal FWD, and therefore prevented from being supplied to the base electrode of the drive transistor Q6. However, the second partial current If2 of the control current Ic1 is allowed to be supplied to the base electrode of the drive transistor Q4 and the second partial current Ir2 of the control current Ic2 is allowed to be supplied to the base electrode of the drive transistor Q3. In this case, the motor current is gradually attenuated to zero when circling in a closed loop composed of the conductive transistors Q3 and Q4 and the motor M, which is referred to as the brake mode.
When the forward rotation command signal FWD is at the low state and the reverse rotation command signal REV is at the HIGH state, the transistor Q11 is turned off and the transistor Q12 is turned on. As a result, the control current Ic1 supplied from the control voltage source Vc1 is zero and the control current Ic2 supplied from the control voltage source Vc2 has a maximum Ic2m, which may be exemplified by a period T5 shown in FIG. 1(b). At this time, the first partial current Ir1 of the control current Ic2 is allowed to be supplied to the base electrode of the drive transistor Q6 because the transistor Q10 which is controlled by the forward rotation command signal FWD is turned off. As a result, the transistors Q2 and Q3 are turned on for allowing a reverse current to flow from the terminal Nb to the terminal Na, thereby operating the motor M in the reverse rotation mode.
Although the conventional motor control circuit 10 effectively controls the various operational modes of the motor M, several problems are inevitably caused in its practical applications. Referring to FIG. 1(b), a curve 11 illustrates the control current Ic1 supplied from the control voltage source Vc1 of the motor control circuit 10 among the operational modes. As clearly identified in the curve 11, the control voltage source Vc1 supplies the same control current Ic1 with the maximum Ic1m regardless of the forward rotation mode and the brake mode. However, as described above, the first partial current If1 of the control current Ic1 in the brake mode is bypassed down to the ground through the conductive transistor Q9 and has nothing to do with controlling the motor drive circuit 1, resulting in a meaningless waste of power. Referring to FIG. 1(b), a curve 12 illustrates the control current Ic2 supplied from the control voltage source Vc2 of the motor control circuit 10 among the operational modes. As clearly identified in the curve 12, the control voltage source Vc2 supplies the same control current Ic2 with the maximum Ic2m regardless of the reverse rotation mode and the brake mode. However, as described above, the first partial current Ir1 of the control current Ic2 in the brake mode is bypassed down to the ground through the conductive transistor Q10 and has nothing to do with controlling the motor drive circuit 1, resulting in a meaningless waste of power.
Referring to FIG. 1(b), a curve 13 illustrates a total control current consumption of the conventional motor control circuit 10 among the operational modes, i.e. a superposition of the currents Ic1 and Ic2 (the curves 11 and 12). As clearly identified in the curve 13, the total control current consumption in the brake mode is twice larger than that in the forward or reverse rotation mode. As described above, the doubling of the total control current consumption is not required by the practical controlling operation and only causes the meaningless waste of power. Therefore, the conventional motor control circuit 10 has a problem with regard to inefficiency and in turn raises operational temperature due to heat generation. A significantly high temperature may cause the circuit malfunction or shut down by thermal protection mechanism.
Furthermore, the practical magnitude of the control current Ic1 changes in accordance with the control voltage source Vc1 since the control current Ic1 is generated by the control voltage source Vc1 through the transistor Q7 and the bias resistor. Similarly, the practical magnitude of the control current Ic2 changes in accordance with the control voltage source Vc2 since the control current Ic2 is generated by the control voltage source Vc2 through the transistor Q8 and the bias resistor. Because the control currents Ic1 and Ic2 directly affects the operation points of the drive transistors Q1 to Q6, the fluctuation of the control voltage sources Vc1 and Vc2 brings about a problem with regard to operational instability of the motor M.